Phased array device and calibration method therefor

ABSTRACT

The calibration method, performed on a phased array device including channel elements coupled in parallel by a transmission line, has the steps of: obtaining channel responses corresponding to the channel elements through the transmission line, and each of the channel responses is obtained when one of the channel elements is turned on, and the rest of the channel elements are turned off; calculating a characteristic value corresponding to the transmission line based on the obtained channel responses of the channel elements; and adjusting a channel parameter of one of the channel elements based on the characteristic value of the transmission line.

CROSS REFERENCE TO RELATED APPLICATIONS

This application claims priority of U.S. Provisional Application No.61/558,661, filed on 11 Nov. 2011, and the entirety of which isincorporated by reference herein.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to electronic circuits, and in particularrelates to a phased-array device and a calibration method therefor.

2. Description of the Related Art

A phased array is an array of antennas in which the relative phases ofthe respective signals feeding the antennas are varied in phases andgains relationship that the effective radiation pattern of the array isreinforced in a desired direction and suppressed in undesireddirections. Corresponding to the each antenna there is a transmitter andreceiver responsible for managing the phases and gains relationship ofthe respective signal feeding in the antenna. The phased array canutilize digital techniques for antenna beamforming, which modify phasesand amplitudes of digital signals by digitally multiplying a complexweight to each antenna feed. The signals from all elements are thencombined digitally in such a way that signals at particular anglesexperience constructive interference while others experience destructiveinterference, giving a number in digital form whose directional responseis a function of the array geometry and the manipulated digital signals,resulting in a controllable beamforming shape. Each antenna elementfeeds a dedicated receiver channel element, providing amplification andselectivity. The frequency and phase responses of the receiver channelelements must track closely, over the full dynamic range of signals tobe handled.

To obtain the full benefit of a digital beamforming array it isnecessary to calibrate out all phase and gain mismatches of the system.The calibration routine contains injecting a test signal into eachreceiving channel element in turn, and measuring the amplitude and phaseof the received signal. Any departures from the desired amplitude andphase can then be compensated using corrected digital weights. Theperformance in the presence of channel element mismatches is criticalparameter of any phased array. These variations result in amplitude andphase mismatches in the radiated/received signals and hence adverselyaffect the beam pattern. The mismatch may arise from the intra-chipvariation inherent to any process technology.

BRIEF SUMMARY OF THE INVENTION

In one aspect of the invention, a calibration method is disclosed,performed on a phased array device comprising a plurality of channelelements coupled in parallel by a transmission line, the methodcomprising: obtaining a plurality of channel responses corresponding tothe channel elements through the transmission line, wherein each of thechannel responses is obtained when one of the channel elements is turnedon, and the rest of the channel elements are turned off; calculating acharacteristic value corresponding to the transmission line based on theobtained channel responses of the channel elements; and adjusting achannel parameter of one of the channel elements based on thecharacteristic value of the transmission line.

In another aspect of the invention, a phased array device is provided,comprising a transmission line, a plurality of channel elements, and acalibration circuit. The plurality of channel elements are coupled inparallel by the transmission line. The calibration circuit is configuredto obtain a plurality of channel responses corresponding to the channelelements through the transmission line, calculate a characteristic valuecorresponding to the transmission line based on the obtained channelresponses of the channel elements, and adjust a channel parameter of oneof the channel elements based on the characteristic value of thetransmission line. Each of the channel responses is obtained when one ofthe channel elements is turned on, and the rest of the channel elementsare turned off.

BRIEF DESCRIPTION OF DRAWINGS

The invention can be more fully understood by reading the subsequentdetailed description and examples with references made to theaccompanying drawings, wherein:

FIG. 1 is a block diagram of a phased-array device according to anembodiment of the invention;

FIG. 2 is a block diagram of a phased-array device according to anotherembodiment of the invention;

FIG. 3 is a flowchart of a gain calibration method for a phased-arrayreceiver according to an embodiment of the invention;

FIG. 4 is a flowchart of a phase calibration method for a phased-arrayreceiver according to an embodiment of the invention;

FIG. 5 is a flowchart of a gain calibration method for a phased-arraytransmitter according to an embodiment of the invention; and

FIG. 6 is a flowchart of a phase calibration method for a phased-arraytransmitter according to an embodiment of the invention.

DETAILED DESCRIPTION OF THE INVENTION

A detailed description is given in the following embodiments withreference to the accompanying drawings.

FIG. 1 is a block diagram of a phased-array device 1 (phased arraydevice) according to an embodiment of the invention, comprising abaseband circuit 100, down-converter and filter 102, modulator 104,channel elements 106 a-d, Loopback Power Amplifier (LBPA) 108, LoopbackLow Noise Amplifier (LBLNA) 110, demodulator 112, up-converter andfilter 114, power meter (PM) 116, switches SW1-SW4, resistors RL1 andRL2, and transmission line L_(TL).

The channel elements 106 a-d can comprise transmitter front ends and/orreceiver front ends. Each channel element may suffer from achannel-to-channel mismatch due to differences in circuit elements andconnection routes in the transmitter front ends and/or receiver frontends. The channel elements 106 a-d are connected in an equal distance 1₂ to one another by the transmission line L_(TL). The transmission lineL_(TL) connects all the channel elements 106 a-d, the loopback PA 108and the loopback LNA 110 in parallel, forming signal loops between thechannel elements 106 a-d and the loopback LNA 110 for detectingchannel-to-channel mismatch for the transmission paths, and between theloopback PA 108 and the channel elements 106 a-d for detectingchannel-to-channel mismatch for the receive paths. Further, channelelements 106 a-d are separated by a substantially equal distance 1 ₂ toone another on the transmission line L_(TL). Only one of thechannel-to-channel mismatches for the transmission paths and the receivepaths may be determined at a time since the channel-to-channel mismatchdetection procedures share the same transmission line L_(TL). Therefore,only one of the loopback LNA 110 and the loopback PA 108 is required tobe activated at a time, for determining the channel-to-channel mismatchfor the transmission paths or receive paths respectively.

Accordingly, when detecting the mismatches in transmission paths, theloopback LNA 110 is turned on while the loopback PA 108 remains off. Themismatch detection procedure is conducted for each of the four channelelements 106 a-d in turn to collect a plurality of channel responses Rkcorresponding to the channel elements 106 a-d through the transmissionline. For example, when taking a mismatch measurement for thetransmitter front end in the channel element 106 a, the channel element106 a is turned on while the channel elements 106 b-d remain off. Thebaseband circuit 100 can inject a test signal St through the channelelement 106 a onto the transmission line L_(TL), on which the loopbackLNA 110 can detect the injected test signal St′ to obtain a channelresponse R0 for determining the channel-to-channel mismatch for thetransmitter front end of the channel element 106 a. The baseband circuit100 can in turn send the test signal St by the remaining channelelements 106 b-d to gather the channel responses Rk for the rest of thechannel elements, with k being 1-3 corresponding to the remainingchannel elements 106 b-d. Specifically in the embodiment provided inFIG. 1, the loopback LNA 110 can collect 4 channel responses R0-R3corresponding to the 4 phased array channel elements 106 a-d. Based onthe 4 channel responses Rk, the baseband circuit 100 can estimate acharacteristic value for the transmission line L_(TL), from which thebaseband circuit 100 can further determine a mismatch parameter foradjusting a channel parameter of the channel elements 106 a-d. Thecharacteristic value may be associated with a transmission lineattenuation constant α or a transmission line phase constant β of thetransmission line L_(TL). The mismatch parameter may be associated witha gain mismatch or a phase mismatch between the channel elements. Thechannel parameter is a control signal for varying amplitude orcontrolling a phase shift of the output signal of transmitter front endof the channel element 106. The test signal St may be orthogonal signalsat baseband.

Similarly, when testing for the mismatches in receive paths, theloopback PA 108 is turned on and the loopback LNA 110 is turned off. Themismatch detection procedure is likewise conducted for each of the fourchannel elements 106 a-d in turn to collect a plurality of channelresponses Rk corresponding to the channel elements 106 a-d through thetransmission line L_(TL). For example, when taking a mismatchmeasurement for the receiver front end in the channel element 106 a, thechannel element 106 a is turned on while the channel elements 106 b-dremain off. The baseband circuit 100 can inject a test signal St throughthe loopback PA 108 onto the transmission line L_(TL), on which thereceiver front end in the channel element 106 can detect the injectedtest signal St′ to obtain the channel response Rk for determining thechannel-to-channel mismatch for the receiver front end of the channelelement 106 a. The baseband circuit 100 can continue sending the testsignal St by loopback PA 108 to collect the channel responses Rk for allchannel elements by the channel elements 106 b-d in turn. Specifically,the channel elements 106 a-d can detect 4 channel responses R0-R3. Basedon the 4 channel responses R0-R3, the baseband circuit 100 can estimatea characteristic value for the transmission line L_(TL), from which thebaseband circuit 100 can further determine a mismatch parameter foradjusting a channel parameter of the channel elements 106 a-d. Thecharacteristic value may be associated with the transmission lineconstant α or β of the transmission line L_(TL). The mismatch parametermay be associated with a gain mismatch or a phase mismatch between thechannel elements. The channel parameter may be a control signal forvarying amplitude or controlling a phase shift of the output signal ofreceiver front end of the channel element 106. The channel parametersdetermined for the receiver front ends may not be the same as thosedetermined for the transmitter front ends. The baseband circuit 100 mayneed to generate separate channel parameters or control signals for thetransmitter and receiver front ends.

The baseband circuit 100 is configured to control all digital signalprocessing, including generating an outgoing baseband signal for themismatch measurement or normal transmission procedure, and processing anincoming baseband signal for the mismatch measurement or normalreception procedure. The baseband circuit 100 comprises a calibrationcircuit 1000, receiving the channel responses Rk from either theloopback LNA 110 or the receiver front ends of the channel elements 10a-d, determining the characteristic values and mismatch parameters basedon the received channel responses Rk, and generating the channelparameters for adjusting the amplitude and phase of the correspondingtransmitter or receiver front end so that the gain and phase mismatchesbetween channels can be calibrated or corrected. In the gain mismatchcalibration, the gains for all element channels 106 a-d are adjusted bydigital gain control signals to render substantially the same amplitudesof output signals for the element channels 106 a-d. In the phasemismatch calibration, the phase shifts for all element channels 106 a-dare adjusted by digital phase control signals to render substantially nophase difference among the output signals of the element channels 106a-d.

The determination procedure of the characteristic values, mismatchparameters and the channel parameters for channel elements 106 a-d aredetailed in gain calibration methods 3 and 5 and phase calibrationmethods 4 and 6. The outgoing baseband signal can undergo a series ofsignal processing including digital to analog conversion, up-conversionand other filtering processes in the DAC, up-converter and filters 114and modulation in the modulator 112 to render the loopback PA 108 or thetransmitter front ends of the channel elements 106 a-d the RF inputsignals. Conversely, the RF input signals coming out of the loopback LNA110 and the receiver front ends of the channel elements 106-d can gothrough demodulation in the demodulator 104 and filtering,down-conversion, and analog-to-digital conversion in the ADC,down-converter and filters 102 to render the channel response Rk for thecalibration circuit 1000 for the mismatch detection and calibrationprocedure.

The loopback PA 108 and the loopback LNA 110 are identical to thecircuits of the transmitter front ends and the receiver front endsrespectively. The power meter 116 is attached to an output of theloopback PA 108, at which a power level of the injected test signal St′is detected by the power meter 116. The power level of the injected testsignal St′ is closely monitored so that the gain of the loopback PA 108can be adjusted to meet a proper received power range. The switches SW1and SW3 can be closed to turn on the loopback LNA 110, and likewise, theswitches SW2 and SW4 can be closed to turn on the loopback PA 108.Conversely, the switches SW1 and SW3 can be opened to turn off theloopback LNA 110, and the switches SW2 and SW4 can be opened to turn offthe loopback PA 108. The on and off states of the switches SW1-SW4 canbe controlled by digital control signals generated from the basebandcircuit 100. The resistor RL1 serves to provide an input matching forthe loopback LAN 110 when being connected to the loopback LNA 110 by theswitch SW1. The resistor RL2 serves to provide an output matching forthe loopback PA 108 when being connected to the loopback PA 108 by theswitch SW2.

The transmitter front end may contain a pre-driver, a transmitter phaseshifter, a power amplifier, and switches. The transmitter phase shiftermay adjust a transmitter phase of a transmitter output signal to accountfor a phase difference mismatch between the transmitter front ends inchannel elements 106 a-d. The transmitter phase shifter may also adjustthe transmitter phase of the transmitter output signal to produce atransmitter output signal with a desired phase shift. In someembodiments, the same transmitter phase shifter is used for the phasemismatch correction and the phase shift generation. In otherembodiments, separate transmitter phase shifters are used for the phasemismatch correction and the phase shift generation. The transmitterphase shifter can adjust the transmitter phase of the transmitter outputsignal according to a digital phase control signal (not shown). Thepower amplifier can increase power of the transmitter input signals witha transmitter gain, adjustable by a digital gain control signal (notshown) to produce the output signal with wanted amplitude. The receiverfront end may comprise a receiver phase shifter, a low noise amplifierand switches. Similar to the transmitter phase shifter, the receiverphase shifter may adjust a phase of a receiver output signal for thereceiver front end to remove or reduce a phase difference mismatchbetween the receiver front ends in channel elements 106 a-d, or, mayadjust the phase of the receiver output signal for the receiver frontend to produce a receiver output signal with a desired phase shift. Thephase mismatch correction and the phase shift generation of the receiveroutput signal may be implemented by a shared or separate receiver phaseshifters. The receiver phase shifter can adjust the receiver phase ofthe receiver output signal according to a digital phase control signal(not shown). The low noise amplifier amplifies power of the receiveroutput signals with a receiver front end gain, adjustable by a digitalgain control signal (not shown) to render wanted amplitude for thereceiver output signal. In some embodiments, the transmitter phaseshifter and the receiver phase shifter are implemented by a single phaseshifter, shared by both the transmitter front end and receiver front endin each channel element. A pair of the transmitter and receiver frontends may share the same antenna and the same signal path to themodulator 112/demodulator 104, and the switch is used to select betweenthe transmit path and the receive path.

FIG. 2 is a block diagram of a phased-array device 2 according toanother embodiment of the invention. The phased-array device 2 is a16-channel elements phased array, similar to the phased-array device 1,except that each channel element in the phased-array device 2 comprisesa transmitter front end TX and a receiver front end RX.

When the phased-array device 2 is in a factory test or being powered-on,the channel-to-channel calibration methods 3-6 can be launched to reduceor remove the gain and phase mismatches among transmitter front endsTX0-TX15, and among receiver front ends RX0-RX15. The interval betweenthe channel-to-channel calibrations will depend on the timescale onwhich the channel responses may be expected to vary. Thechannel-to-channel calibration methods are applicable only to thechannel elements spaced by the equal distance 1 ₂ on the transmissionline L_(TL). For example, the transmitter front ends TX0-TX7 are groupedinto one group, and the transmitter front ends TX8-TX15 are grouped intoanother group for performing the channel-to-channel calibration methods3-6. The more elements in one group, the more accurate the gain or phasemismatch can be estimated and removed. When all transmitter front endsRX0-RX15 are arranged in a loop, separated by the equal distance 1 ₂ toone another, there is no error in the gain or phase mismatch estimationby using the channel-to-channel calibration methods 3-6.

FIG. 3 is a flowchart of a channel-to-channel gain calibration method 3for the receive paths according to an embodiment of the invention,incorporating the phased array devices 1 or 2 in the invention.

Upon startup of the gain calibration method 3, the phased-array device 1is configured to activate the loopback PA 108, deactivate the loopbackLNA 110, and prepare the loopback PA 108 for initiating the test signalSt on the transmission line L_(TL) (S300). Further, the phased-arraydevice 1 is configured to turn only on the first channel element, suchas the channel 106 a, while keeping the rest of the channel elements(channel 106 b-d) off (S302), so that the channel 106 a is the only onechannel that can receive the signals from the transmission line L_(TL).The receiver front end in the channel 106 a is set to a default digitalgain setting D_(G0) and a default digital phase setting D_(φ0). Theloopback PA 108 can place the test signal St′ on the transmission lineL_(TL) for the receiver front end to pick up (S304). In response, thefirst channel 106 a can acquire the test signal St', down-convert anddigitize the test signal St′ to derive a first receiver channel responseR0 for the calibration circuit 1000 to determine the channel-to-channelmismatch (S306). The first receiver channel response R0 can be expressedas S_(BB0)exp(jθ_(BB0)), with SBB0 being the amplitude and θ_(BB0) beingthe phase shift of the channel response signal R0 with reference to thetest signal St.

The phased-array device 1 can then turn on only the second channelelement 106 b and turn off the remaining channels 106 a, c-d (S308),send the test signal St′ again onto the transmission line L_(TL) throughthe loopback PA 108 (S310), and obtain the test signal St′ through thereceiver front end of the second channel element 106 b (S312). Whileacquiring the test signal St′ from the transmission line L_(TL), thereceiver front end of the channel 106 b is set to a default digital gainsetting DG1 and a default digital phase setting Dφ1. The test signal St′undergoes every circuit element and process on the receive path via thesecond channel element 106 b to be the second channel response R1 forthe calibration circuit 1000 to determine the channel-to-channelmismatch. The second receiver channel response R1 can be expressed asS_(BB1)exp(jθ_(BB1)), with SBB1 being the amplitude and θ_(BB1) beingthe phase shift of the channel response signal R1 with reference to thetest signal St. The channel-to-channel gain calibration method 3 onlyturn on one channel element 106 at a time, to measure the correspondingchannel response specifically for the turned-on channel.

Next, the gain calibration method 3 continues to switch the activechannel for taking the next channel response Rk, with k being thechannel count 0-3 in the case of phased-array device 1. The calibrationcircuit 1000 or a channel count circuit (not shown) can determinewhether all channels in the phased-array device 1 have the channelresponse measurements SBBK taken for (S314).

When all channel responses RK are collected, the calibration circuit1000 can now calculate characteristic value exp(−α1 ₂) for thetransmission line L_(TL) according to the amplitudes S_(BBk) of thechannel responses RK (S316). Specifically, the calibration circuit 1000can compute a gain mismatch parameter

$\frac{S_{BBk}}{S_{{BB}{({k + 1})}}}$

between any two adjacent channel elements 106, as expressed by Eq. (1):

$\begin{matrix}{{\Delta \; S_{BBk}} = {\frac{S_{BBk}}{S_{{BB}{({k + 1})}}} = {{\exp \left( {{- \alpha}\; l_{2}} \right)}\frac{G_{k}}{G_{k + 1}}}}} & {{Eq}.\mspace{14mu} (1)}\end{matrix}$

where G_(k) is a current gain of a current channel element 106;

G_(k+1) is a next gain of a next channel element 106; and

exp(−α1 ₂) represents a transmission line gain (or attenuation) due tothe transmission line length 1 ₂.

The channel-to-channel gain mismatch calibration is used to calibratethe channel gain G_(k) for all channel element 106 a-d, so that allchannel gains G_(k) are substantially the same to one another, or gainmismatch G_(k)/G_(k+1)≈1. Therefore the calibration circuit 1000 candetermine the gain mismatch parameter ΔS_(BBk) for all adjacent channelelements, then determine the characteristic value exp(−α1 ₂) for thetransmission line L_(TL) based on all gain mismatch parameters, asexpressed by Eq. (2) and Eq. (3):

$\begin{matrix}{{\Delta \; S_{{BB}\; 1} \times \Delta \; S_{{BB}\; 2}\mspace{14mu} \ldots \times \Delta \; S_{BBn}} = {{\exp \left( {{- n}\; \alpha \; l_{2}} \right)}\frac{G_{0}}{G_{n}}}} & {{Eq}.\mspace{14mu} (2)}\end{matrix}$

where (n+1) is a number of the channel elements 106 separated by length1 ₂.

$\begin{matrix}{\left. \Rightarrow{\exp \left( {{- \alpha}\; l_{2}} \right)} \right. = \sqrt[n]{\frac{\Delta \; S_{{BB}\; 1} \times \Delta \; S_{{BB}\; 2}\ldots \times \Delta \; S_{BBn}}{\frac{G_{0}}{G_{n}}}}} & {{Eq}.\mspace{14mu} (3)}\end{matrix}$

Eq. (3) shows that the characteristic value exp(−α1 ₂) for thetransmission line L_(TL) only depends on the gain mismatch parametersΔS_(BBk) and (G₀/G_(n)). Eq. (3) also indicates that the characteristicvalue exp(−α1 ₂) can be determined based on a product of the gainmismatch parameters

$\frac{S_{BBk}}{S_{{BB}{({k + 1})}}}$

(characteristic differences). The term (G₀/G_(n)) is kept within 1 dBerror by circuit design or circuit configuration, therefore the gainmismatch (G_(k)/G_(k+1)) can be kept within 1/n dB error. The error ofthe determined characteristic value exp(−α1 ₂) decreases with anincrease in the number n of the channel elements 106 separated by length1 ₂. It follows that the accuracy of the estimated gain mismatch(G_(k)/G_(k+1)) increases with the number n of the channel elements 106.It is noted that in some embodiments, all channel elements 106 areplaced in a loop, each spaced by the transmission line length 1 ₂ to theadjacent channel element 106, the gain G_(n) for the last channelelement 106 can be considered as the gain G₀ for the first channelelement 106, and the term (G₀/G_(n)) is reduced to 1, or 0 dB error,consequently the estimated gain mismatch (G_(k)/G_(k+1)) can also bekept at 0 dB error.

The gain calibration method 3 can determine the gain mismatch(G_(k)/G_(k+1)) according to Eq. (4).

$\begin{matrix}{\left. \Rightarrow\frac{G_{k}}{G_{k + 1}} \right. = \frac{\Delta \; S_{BBk}}{\sqrt[n]{\frac{\Delta \; S_{{BB}\; 1} \times \Delta \; S_{{BB}\; 2}\ldots \times \Delta \; S_{BBn}}{\frac{G_{0}}{G_{n}}}}}} & {{Eq}.\mspace{14mu} (4)}\end{matrix}$

The calibration circuit 1000 can adjust the receiver gain of the secondchannel element 106 b with reference to the receiver gain of the firstchannel element 106 a (S318), then run Steps S308-S312 again fordetermining the channel response R1′ of the second channel element 106b. Accordingly, the receiver gains of the receiver front ends can beadjusted by the digital gain control signals. Based on the new channelresponse R1′, the calibration circuit 1000 can determine whether(S_(BB1)′/S_(BB0)) now approaches to the characteristic value exp(−α1 ₂)within a predetermined range set during the circuit design. Thecalibration circuit 1000 can adjust the receiver gain again until(S_(BB1)′/S_(BB0)) is substantially the same as the characteristic valueexp(−α1 ₂).

Following by the gain adjustment for the second channel element, thecalibration circuit 1000 next can adjust the receiver gain for the nextchannel element 106 c with reference to the receiver gain of theprevious channel element 106 b and determine the new channel responseR2′ after the gain calibration. Based on the newly determined channelresponse R2′, the calibration circuit 1000 can further determine whetherthe gain mismatch parameter (S_(BB2)′/S_(BB1)) is substantially the sameas the characteristic value exp(−α1₂). If so, the gain calibration forthe third receiver front end is then completed, and if not, thecalibration circuit 1000 can continue adjusting the receiver gain of thethird channel element 106 c until the (S_(BB2)′/S_(BB1)) approaches thecharacteristic value exp(−α1 ₂) to the predetermined range.

The calibration circuit 1000 can perform the gain adjustment procedurefor the remaining channel element 106 d. The gain calibration method 3is completed and exited after all receiver front ends of the channelelements 106 have been calibrated for the channel-to-channel gainmismatch.

Taking the phased-array device 2 as an example, to determine the gainmismatches for the receiver front ends RX0-RX7, the gain calibrationmethod 3 is implemented to turn on the loopback PA 108 and turn off theloopback LNA 110 (S300), turn the first receiver front end RX0 on andthe remaining receiver front ends RX1-RX7 off (S302), initiate the testsignal St′ on the transmission line LTL by loopback PA 108 (S304),obtain the first channel response R0 by the first receiver front end RX0(S306), hence complete the round for determining the channel response R0for the first receiver front end RX0. The gain calibration method 3 thencarries on to obtain the other 7 channel responses R1-R7 for theremaining receiver front ends RX1-RX7 based on the loop outlined in StepS308-S314 until the channel responses for all receiver front endsRX0-RX7 are collected. The gain calibration method 3 next is implementedto calculate the characteristic value exp(−α1 ₂) by applying the channelresponses R0-R7 into Eq. (1), (2), and (3) (S316), and adjust gainsettings for the receiver front ends RX1-RX7 by the digital gain controlsignals until all gain mismatch parameters

$\frac{S_{{BB}\; 0}}{S_{{BB}\; 1}},\frac{S_{{BB}\; 1}}{S_{{BB}\; 2}},\ldots \mspace{14mu},\frac{S_{{BB}\; 6}}{S_{{BB}\; 7}}$

equal to the characteristic value exp(−α1 ₂) (S318).

The gain calibration method 3 employs a plurality of channel responsesRk for computing the characteristic value exp(−α1 ₂), thereby reducingthe error in estimated characteristic value exp(−α1 ₂) and increasingthe accuracy for determining the channel-to-channel gain mismatch.Further, when all channel elements 106 are placed in a loop with anequal distance 1 ₂ to the adjacent channel elements 106, the error forgain mismatch determination can be reduced to 0.

FIG. 4 is a flowchart of a channel-to-channel phase calibration method 4for the receive paths according to an embodiment of the invention,incorporating the phased array devices 1 or 2 in the invention.

Steps S400 through S414 are identical to Steps S300 through S314,reference therefor can be found in the preceding paragraph and will notbe repeated here for brevity. Accordingly, the channel response R1 forthe first channel element 106 a can be expressed byS_(BB1)exp(jθ_(BB1)), the channel response R2 for the second channelelement 106 b can be expressed by S_(BB2)exp(jθ_(BB2)). For thechannel-to-channel phase calibration, the calibration circuit 1000reduces or removes the phase difference between channel elements, i.e.,θ_(BBk)−θ_(BB(k+1))≈0.

After all channel responses RK are collected, the calibration circuit1000 can now calculate characteristic value (β1 ₂) for the transmissionline L_(TL) according to the phase θ_(BBk) of the channel response RK(S416). Specifically, the calibration circuit 1000 can compute a phasemismatch parameter Δθ_(BBk) between any two adjacent channel elements106, as expressed by Eq. (5):

In step S416

Δθ_(BBk)=θ_(BBk)−θ_(BB(k+1)=−β)1₂+θ_(k)−θ_(k+1)  Eq. (5)

where θ_(BBk) is a current phase value of a current channel element 106;

θ_(BBk+1) is a next phase value of a next channel element 106; and

β1 ₂ represents a phase shift due to the transmission line length 1 ₂.

The calibration circuit 1000 can determine the phase mismatch parameterΔθ_(BBk) for all adjacent channel elements, then determine thecharacteristic value (β1 ₂) for the transmission line L_(TL) based onall phase mismatch parameters Δθ_(BBk), as expressed by Eq. (6) and Eq.(7):

$\begin{matrix}{{\sum\limits_{k = 1}^{n - 1}\; {\Delta\theta}_{BBk}} = {{{- \left( {n - 1} \right)}\beta \; l_{2}} + \theta_{1} - \theta_{n}}} & {{Eq}.\mspace{14mu} (6)}\end{matrix}$

where n is a number of the channel elements 106 separated by length 1 ₂.

$\begin{matrix}{{\beta \; l_{2}} = \frac{\theta_{1} - \theta_{n} - {\sum\limits_{k = 1}^{n - 1}\; {\Delta \; \theta_{BBk}}}}{n - 1}} & {{Eq}.\mspace{14mu} (7)}\end{matrix}$

Eq. (7) shows that the characteristic value (β1 ₂) for the transmissionline L_(TL) only depends on the phase mismatch parameter Δθ_(BBk) and(θ₁-θ_(n)). Further, Eq. (7) also indicates that the characteristicvalue (β1 ₂) may be determined based on a sum of the phase mismatchparameter Δθ_(BBk) (characteristic differences). The phase mismatchparameter Δθ_(BBk) is known by the phases θ_(k) and θ_(k+1). The term(θ₁-θ_(n)) can be kept within 10 degree error by circuit design orcircuit configuration, therefore the error of the estimated phasemismatch (θ_(k)-θ_(k+1)) can be kept within 10/(n−1) degree. It can beseen in Eq. (7) that the error of the determined characteristic value(β1 ₂) decreases with an increase in the number n of the channelelements 106 separated by length 1 ₂. It follows that the accuracy ofthe estimated phase mismatch (θ_(k)-θ_(k+1)) increases with the number nof the channel elements 106. In some embodiments, all channel elements106 are placed in a loop, each spaced by the transmission line length 1₂ to the adjacent channel element 106, the phase θ_(n) for the lastchannel element 106 can be considered as the phase θ₁ for the firstchannel element 106, and the term (θ₁-θ_(n)) can be reduced to 0,consequently the estimated phase mismatch (θ_(k)-θ_(k+1)) can also bekept at 0 degree.

The phase calibration method 4 can determine the phase mismatch(θ_(k)-θ_(k+1)) according to Eq. (8).

$\begin{matrix}{{\theta_{k} - \theta_{k + 1}} = {{{\Delta \; \theta_{BBk}} + {\beta \; l_{2}}} = {{\Delta \; \theta_{BBk}} + \frac{\theta_{1} - \theta_{n} - {\sum\limits_{k = 1}^{n - 1}\; {\Delta \; \theta_{BBk}}}}{n - 1}}}} & {{Eq}.\mspace{14mu} (8)}\end{matrix}$

The calibration circuit 1000 can adjust the receiver phase of the secondchannel element 106 b with reference to the receiver phase of the firstchannel element 106 a (S418), then run Steps S408-S412 again fordetermining an updated channel response R1′ of the second channelelement 106 b. Accordingly, the receiver phases of the receiver frontends can be adjusted by the digital phase control signals. Based on theupdated channel response R1′, the calibration circuit 1000 can determinewhether (θ_(BB0)-θ_(BB1)′) approaches to the characteristic value (β1 ₂)within a predetermined range set during the circuit design. Thecalibration circuit 1000 can adjust the receiver phase until(θ_(BB0)-θ_(BB1)′) is substantially the same as the characteristic value(β1 ₂).

Following by the phase adjustment for the second channel element, thecalibration circuit 1000 next can adjust the receiver phase for the nextchannel element 106 c with reference to the receiver phase of theprevious channel element 106 b and determine the new channel responseR2′ after the phase calibration. Based on the newly determined channelresponse R2′, the calibration circuit 1000 can further determine whetherthe phase mismatch parameter (θ_(BB1)-θ_(BB2)′) is substantially thesame as the characteristic value (β1 ₂). If so, the phase calibrationfor the third receiver front end is then completed, and if not, thecalibration circuit 1000 can continue adjusting the receiver phase ofthe third channel element 106 c until the (θ_(BB1)-θ_(BB2)′) approachesthe characteristic value (β1 ₂) to the predetermined range.

The calibration circuit 1000 can perform the phase adjustment procedurefor the remaining channel element 106 d. The phase calibration method 4is completed and exited after all receiver front ends of the channelelements 106 have been calibrated for the channel-to-channel phasemismatch.

Taking the phased-array device 2 as an example, to determine the phasemismatches for the receiver front ends RX0-RX7, the phase calibrationmethod 4 is implemented to turn on the loopback PA 108 and turn off theloopback LNA 110 (S400), turn the first receiver front end RX0 on andthe remaining receiver front ends RX1-RX7 off (S402), initiate the testsignal St′ on the transmission line LTL by loopback PA 108 (S404),obtain the first channel response R0 by the first receiver front end RX0(S406), hence complete the round for determining the channel response R0for the first receiver front end RX0. The phase calibration method 4then carries on to obtain the other 7 channel responses R1-R7 for theremaining receiver front ends RX1-RX7 based on the loop outlined in StepS408-S414 until the channel responses for all receiver front endsRX0-RX7 are collected. The phase calibration method 4 next isimplemented to calculate the characteristic value (β1 ₂) by applying thechannel responses R0-R7 into Eq. (5), (6), and (7) (S416), and adjustphase settings for the receiver front ends RX1-RX7 by the digital phasecontrol signals until all phase mismatch parameters (θ_(BB0)-θ_(BB1)),(θ_(BB1)-θ_(BB2)), . . . , (θ_(BB7)-θ_(BB6)) equal to the characteristicvalue exp(β1 ₂) (S418).

The phase calibration method 4 employs a plurality of channel responsesRk for computing the characteristic value (β1 ₂) of the transmissionline segment 1 ₂, thereby reducing the error in the estimatedcharacteristic value (β1 ₂) and increasing the accuracy for determiningthe channel-to-channel phase mismatch (θ_(k)-θ_(k+1)). Further, when allchannel elements 106 are placed in a loop with an equal distance 1 ₂ tothe adjacent channel elements 106, the error for phase mismatchdetermination can be reduced to 0.

FIG. 5 is a flowchart of a channel-to-channel gain calibration method 5for the transmission paths according to an embodiment of the invention,incorporating the phased array devices 1 or 2 in the invention.

Upon startup of the gain calibration method 5, the phased-array device 1is configured to activate the loopback LNA 110, deactivate the loopbackPA 108, and prepare the loopback LNA 110 for initiating the test signalSt on the transmission line L_(TL) (S500). Further, the phased-arraydevice 1 is configured to turn only on the first transmitter front endof the channel element, such as the channel element 106 a, while keepingthe transmitter front ends the rest of the channel elements (channel 106b-d) off (S502), so that the channel 106 a is the only one channel thatcan transmit the signals onto the transmission line L_(TL). Thetransmitter front end in the channel 106 a is set to a default digitalgain setting D_(G0) and a default digital phase setting D_(φ0). Thefirst channel element 106 a can send the test signal St′ onto thetransmission line L_(TL) for the loopback LNA 110 to pick up (S504). Inresponse, the loopback LNA 110 can acquire the test signal St',down-convert and digitize the test signal St′ to derive a firsttransmitter channel response R0 for the calibration circuit 1000 todetermine the channel-to-channel mismatch (S506). The first transmitterchannel response R0 can be expressed as S_(BB0)exp(jθ_(BB0)), with SBB0being the amplitude and θ_(BB0) being the phase shift of the channelresponse signal R0 with reference to the test signal St.

The phased-array device 1 can then turn on only the second transmitterfront end of the channel element 106 b and turn off the transmitterfront ends of the remaining channel elements 106 a, c-d (S508), send thetest signal St′ again onto the transmission line L_(TL) through thesecond transmitter front end (S510), and obtain the test signal St′through the loopback LNA 110 (S512). While delivering the test signalSt′ onto the transmission line L_(TL), the transmitter front end of thechannel 106 b is set to a default digital gain setting D_(G1) and adefault digital phase setting D_(φ1). The test signal St′ undergoesevery circuit element and process on the transmission path via thesecond channel element 106 b and loops back to the baseband circuit 100via the loopback LNA 110 to be the second channel response R2 fordetermining the channel-to-channel transmitter gain mismatch. The secondtransmitter channel response R1 can be expressed asS_(BB1)exp(jθ_(BB1)), with SBB1 being the amplitude and θ_(BB1) beingthe phase shift of the channel response signal R1 with reference to thetest signal St. The channel-to-channel gain calibration method 5 onlyturn on one channel element 106 at a time, to measure the correspondingchannel response specifically for the turned-on channel.

Next, the gain calibration method 5 continues to switch the activechannel for taking the next channel response Rk, with k being thechannel count 0-3 in the case of phased-array device 1. The calibrationcircuit 1000 or a channel count circuit (not shown) can determinewhether all channels in the phased-array device 1 have the channelresponse measurements S_(BBK) taken (S514). When all channel responsesRK are collected, the calibration circuit 1000 can calculatecharacteristic value exp(−α1 ₂) for the transmission line L_(TL)according to the amplitudes S_(BBk) of the channel responses RK (S516).The calibration circuit 1000 can compute a gain mismatch parameterΔS_(BBk) between any two adjacent channel elements 106 according to Eq.(1). After the gain mismatch parameters ΔS_(BBk) for all adjacentchannel elements are computed, the calibration circuit 1000 candetermine the characteristic value exp(−α1 ₂) for the transmission lineL_(TL) based on all gain mismatch parameters ΔS_(BBk) according to Eq.(2) and Eq. (3), and determine the gain mismatch (G_(k)/G_(k+1))according to Eq. (4), and adjust the gain settings for the transmitterfront ends of the channel elements 106 b-d according to the estimatedgain mismatch (G_(k)/G_(k+1)) by the gain adjustment procedure outlinedin the gain calibration method 3. Steps S516 to S518 in FIG. 5 areidentical to Step S316-S318 in FIG. 3, relevant explanation for the gainadjustment can be found in the preceding paragraphs. The gaincalibration method 5 is completed and exited after the transmitter frontends of the channel elements 106 b-d have been calibrated for thechannel-to-channel gain mismatch.

Taking the phased-array device 2 as an example, to determine the gainmismatches for the transmitter front ends TX0-TX7, the gain calibrationmethod 5 is implemented to turn on the loopback LNA 110 and turn off theloopback PA 108 (S500), turn the first transmitter front end TX0 on andthe remaining transmitter front ends TX1-TX7 off (S502), place the testsignal St′ on the transmission line L_(TL) by the first transmitterfront end TX0 (S504), obtain the first channel response R0 by theloopback LNA 110 (S506), hence complete the round for determining thechannel response R0 for the first transmitter front end TX0. The gaincalibration method 5 then carries on to obtain the other 7 channelresponses R1-R7 for the remaining transmitter front ends TX1-TX7 basedon the loop outlined in Step S508-S514 until the channel responses forall transmitter front ends TX0-TX7 are collected. The gain calibrationmethod 5 next is implemented to calculate the characteristic valueexp(−α1 ₂) by applying the channel responses R0-R7 into Eq. (1), (2),and (3) (S516), and adjust gain settings for the transmitter front endsTX1-TX7 by the digital gain control signals until all gain mismatchparameters

$\frac{S_{{BB}\; 0}}{S_{{BB}\; 1}},\frac{S_{{BB}\; 1}}{S_{{BB}\; 2}},\ldots \mspace{14mu},\frac{S_{{BB}\; 6}}{S_{{BB}\; 7}}$

equal to the characteristic value exp(−α1 ₂) (S518).

The gain calibration method 5 employs a plurality of channel responsesRk for computing the characteristic value exp(−α1 ₂) of the transmissionline segment 1 ₂, thereby reducing the error in estimated characteristicvalue exp(−α1 ₂) and increasing the accuracy for determining thechannel-to-channel gain mismatch. Further, when all channel elements 106are placed in a loop with an equal distance 1 ₂ to the adjacent channelelements 106, the error for gain mismatch determination can be reducedto 0.

FIG. 6 is a flowchart of a channel-to-channel phase calibration method 6for the transmission paths according to an embodiment of the invention,incorporating the phased array devices 1 or 2 in the invention.

Steps S600 through S614 are identical to Steps S500 through S514 in FIG.5, Steps S616 through S618 are identical to Steps S416 through S418 inFIG. 4, reference therefor can be found in the preceding paragraph andwill not be repeated here for brevity. Accordingly, the channel responseR1 for the first channel element 106 a can be expressed byS_(BB1)exp(jθ_(BB1)), the channel response R2 for the second channelelement 106 b can be expressed by S_(BB2)exp(jθ_(BB2)). For thechannel-to-channel phase calibration, the calibration circuit 1000serves to reduce or remove the phase difference between the transmitterfront ends of the channel elements 106, i.e., θ_(BBk)−θ_(BB(k+1))≈0.

Taking the phased-array device 2 as an example, to determine the phasemismatches for the transmitter front ends TX0-TX7, the phase calibrationmethod 6 is implemented to turn on the loopback LNA 110 and turn off theloopback PA 108 (S600), turn the first transmitter front end TX0 on andthe remaining transmitter front ends TX1-TX7 off (S602), place the testsignal St′ on the transmission line L_(TL) by the first transmitterfront end TX0 (S604), obtain the first channel response R0 by theloopback LNA 110 (S606), hence complete the round for determining thechannel response R0 for the first transmitter front end TX0. The phasecalibration method 6 then carries on to obtain the other 7 channelresponses R1-R7 for the remaining transmitter front ends TX1-TX7 basedon the loop outlined in Step S608-S614 until the channel responses forall transmitter front ends TX0-TX7 are collected. The phase calibrationmethod 6 next is implemented to calculate the characteristic value (β1₂) by applying the channel responses R0-R7 into Eq. (5), (6), and (7)(S616), and adjust phase settings for the transmitter front ends TX1-TX7by the digital phase control signals until all phase mismatch parameters(θ_(BB0)-θ_(BB1)), (θ_(BB1)-θ_(BB2)), . . . , (θ_(BB7)-θ_(BB6)) equal tothe characteristic value exp(β1 ₂) (S618).

The phase calibration method 6 employs a plurality of channel responsesRk for computing the characteristic value (β1 ₂) of the transmissionline segment 1 ₂, thereby reducing the error in the estimatedcharacteristic value (β1 ₂) and increasing the accuracy for determiningthe channel-to-channel phase mismatch (θ_(k)-θ_(k+1)) for thetransmitter front ends. Further, when all channel elements 106 areplaced in a loop with an equal distance 1 ₂ to the adjacent channelelements 106, the error for phase mismatch determination can be reducedto 0.

As used herein, the term “determining” encompasses calculating,computing, processing, deriving, investigating, looking up (e.g.,looking up in a table, a database or another data structure),ascertaining and the like. Also, “determining” may include resolving,selecting, choosing, establishing and the like.

The various illustrative logical blocks, modules and circuits describedin connection with the present disclosure may be implemented orperformed with a general purpose processor, a digital signal processor(DSP), an application specific integrated circuit (ASIC), a fieldprogrammable gate array signal (FPGA) or other programmable logicdevice, discrete gate or transistor logic, discrete hardware componentsor any combination thereof designed to perform the functions describedherein. A general purpose processor may be a microprocessor, but in thealternative, the processor may be any commercially available processor,controller, microcontroller or state machine.

The operations and functions of the various logical blocks, modules, andcircuits described herein may be implemented in circuit hardware orembedded software codes that can be accessed and executed by aprocessor.

While the invention has been described by way of example and in terms ofthe preferred embodiments, it is to be understood that the invention isnot limited to the disclosed embodiments. To the contrary, it isintended to cover various modifications and similar arrangements (aswould be apparent to those skilled in the art). Therefore, the scope ofthe appended claims should be accorded the broadest interpretation so asto encompass all such modifications and similar arrangements.

What is claimed is:
 1. A calibration method, performed on a phased arraydevice comprising a plurality of channel elements coupled in parallel bya transmission line, the method comprising: obtaining a plurality ofchannel responses corresponding to the channel elements through thetransmission line, wherein each of the channel responses is obtainedwhen one of the channel elements is turned on, and the rest of thechannel elements are turned off; calculating a characteristic valuecorresponding to the transmission line based on the obtained channelresponses of the channel elements; and adjusting a channel parameter ofone of the channel elements based on the characteristic value of thetransmission line.
 2. The calibration method of claim 1, wherein thechannel elements are RF transmitter frond-ends or RF receiverfront-ends.
 3. The calibration method of claim 1, wherein when thechannel elements are RF transmitter frond-ends, the transmission line iscoupled to output ports of the RF transmitter frond-ends; when thechannel elements are RF receiver frond-ends, the transmission line iscoupled to input ports of the RF receiver frond-ends.
 4. The calibrationmethod of claim 1, wherein each two adjacent channel elements of thechannel elements are spaced by substantially a same length.
 5. Thecalibration method of claim 1, wherein when the channel elements are RFtransmitter frond-ends, the step of obtaining the plurality of channelresponses corresponding to the channel elements through the transmissionline comprises: sending a test signal to the turned-on RF transmitterfrond-end when the rest of the RF transmitter frond-ends are turned off;receiving a signal outputted by the turned-on RF transmitter frond-endthrough the transmission line; and determining the channel responsecorresponding to the turned-on RF transmitter frond-end based on thereceived signal.
 6. The calibration method of claim 1, wherein when thechannel elements are RF receiver frond-ends, the step of obtaining theplurality of channel responses corresponding to the channel elementsthrough the transmission line comprises: sending a test signal to theturned-on RF receiver frond-end through the transmission line when therest of the RF receiver frond-ends are turned off; receiving a signaloutputted by the turned-on RF receiver frond-end; and determining thechannel response corresponding to the turned-on RF receiver frond-endbased on the received signal.
 7. The calibration method of claim 1,wherein the channel responses corresponding to the channel elements areamplitude values or phase values of signals outputted by the channelelements.
 8. The calibration method of claim 1, wherein the steps ofcalculating the characteristic value corresponding to the transmissionline based on the obtained channel responses of the channel elementscomprises: obtaining a plurality of characteristic differences betweeneach two of the channel responses; determining the characteristic valuecorresponding to the transmission line by the characteristicdifferences.
 9. The calibration method of claim 8, wherein the channelresponses comprises amplitude values outputted by the channel elements,and the step of calculating the characteristic value corresponding tothe transmission line based on the obtained channel responses of thechannel elements comprises: determining the characteristic value basedon a product of the characteristic differences.
 10. The calibrationmethod of claim 8, wherein the channel responses comprises phase valuesoutputted by the channel elements, and the step of calculating thecharacteristic value corresponding to the transmission line based on theobtained channel responses of the channel elements comprises:determining the characteristic value based on a sum of thecharacteristic differences.
 11. The calibration method of claim 1,wherein when the channel responses corresponding to the channel elementsare amplitude values of signals outputted by the channel elements, theadjusted channel parameter is a gain setting of the correspondingchannel element; when the channel responses corresponding to the channelelements are phase values of signals outputted by the channel elements,the adjusted channel parameter is a phase setting of the correspondingchannel element.
 12. A phased array device, comprising: a transmissionline; a plurality of channel elements coupled in parallel by thetransmission line; and a calibration circuit, configured to obtain aplurality of channel responses corresponding to the channel elementsthrough the transmission line, calculate a characteristic valuecorresponding to the transmission line based on the obtained channelresponses of the channel elements, and adjust a channel parameter of oneof the channel elements based on the characteristic value of thetransmission line; wherein each of the channel responses is obtainedwhen one of the channel elements is turned on, and the rest of thechannel elements are turned off.
 13. The phased array device of claim12, wherein when the channel elements are RF transmitter frond-ends, thetransmission line is coupled to output ports of the RF transmitterfrond-ends; when the channel elements are RF receiver frond-ends, thetransmission line is coupled to input ports of the RF receiverfrond-ends.
 14. The phased array device of claim 12, wherein when thechannel elements are RF transmitter frond-ends, the calibration circuitis configured to send a test signal to the turned-on RF transmitterfrond-end when the rest of the RF transmitter frond-ends are turned off,receive a signal outputted by the turned-on RF transmitter frond-endthrough the transmission line, and determine the channel responsecorresponding to the turned-on RF transmitter frond-end based on thereceived signal.
 15. The phased array device of claim 12, wherein whenthe channel elements are RF receiver frond-ends, the calibration circuitis configured to send a test signal to the turned-on RF receiverfrond-end through the transmission line when the rest of the RF receiverfrond-ends are turned off, receive a signal outputted by the turned-onRF receiver frond-end; and determine the channel response correspondingto the turned-on RF receiver frond-end based on the received signal. 16.The phased array device of claim 12, wherein the calibration circuit isconfigured to: obtain a plurality of characteristic differences betweeneach two of the channel responses; and determine the characteristicvalue corresponding to the transmission line by the characteristicdifferences.
 17. The phased array device of claim of claim 15, whereinthe channel responses comprises amplitude values outputted by thechannel elements, and the calibration circuit is configured to determinethe characteristic value based on a product of the characteristicdifferences.
 18. The phased array device of claim of claim 15, whereinthe channel responses comprises phase values outputted by the channelelements, and the calibration circuit is configured to determine thecharacteristic value based on a sum of the characteristic differences.19. The phased array device of claim 12, wherein when the channelresponses corresponding to the channel elements are amplitude values ofsignals outputted by the channel elements, the adjusted channelparameter is a gain setting of the corresponding channel element; whenthe channel responses corresponding to the channel elements are phasevalues of signals outputted by the channel elements, the adjustedchannel parameter is a phase setting of the corresponding channelelement.